Wideband Harmonic Compensation with a Voltage-Source Hybrid Active Power Filter
Active In SP
Joined: Mar 2010
01-04-2010, 03:30 PM
In this seminar and presentation a method for wideband harmonic compensation is presented. The main circuit configuration of the system is shown in Fig. 2. The system compensates the low order harmonics using a simple computational control delay compensation method, while the high order harmonics are filtered with a small passive high pass filter. In the system the whole filtering band of the APF is utilized and the filtering performance is recovered with an HPF when the ability of the APF to filter harmonics effectively is impaired. This way the size of the passive filter can be kept small. First the proposed control system is studied. Then the prototype built and the measurement results at various operating points are presented. The results are compared with other filtering methods. Finally power losses in the hybrid filter system are studied and compared to the losses of the voltage-source shunt APF.
Figure 1. Configuration of the shunt APF
Figure 2. Configuration of the hybrid filter
Figure 3. Block diagram of the control system
II. CONTROL SYSTEM
Fig. 3 presents a block diagram of the proposed control system. Subscripts s, l, f and hp refer to supply; load, active filter and high-pass filter variables respectively, h to harmonics and 0 to fundamental frequency quantities in synchronous reference frame. Underlined variables refer to space vectors, the superscript s to a space vector in synchronously rotating reference frame and * to reference value. The control system is implemented in the synchronously rotating reference frame, which is tied to the supply voltage vector. The task of the system is to produce a filter current if opposite to the harmonics sensed. The measured load and supply currents, il and is respectively, are first transformed into the synchronously rotating reference frame. The reference frame angle is determined with a phase locked loop (PLL) by observing supply voltage us. Next, the harmonics is lh and is sh are extracted from the fundamental current components with a system based on high-pass filtering (blocks HP in Fig. 3). The filter current harmonic reference i*sfh is produced with the help of the feedforward of the load currents il and feedback of the supply currents is. The feedforward observes harmonics generated by the load and the feedback the harmonics that persist in the supply current after active and passive filtering. The feedback also damps the harmonics drawn by the HPF in the case of distorted supply voltages and prevents resonance. The effect of the control delay is compensated with the block CDC in Fig. 3. Since we know the performance of the filter current in case of step change in the current reference, the reference can be corrected so that the filter current behaves as desired. The algorithm can be written as
I*sfhl = [-tc (islh(k)-islh(k-1))]-islh(k)
where Tc is a compensation time constant and Ts sample time. The APF dc link voltage is controlled with a fundamental frequency d-axis current reference component i*fd0. This is a dc quantity in synchronous reference frame and is added to the harmonic reference i*fh. A nonlinear Pe2 controller is used in voltage control. The controller output is proportional to the square of the input . The reactive power produced by the passive filter is compensated with a fundamental frequency q axis reference i*fq0. A PID controller is used in the closed-loop control of the filter currents i*f. Finally the active filter voltage reference u*sf is produced by subtracting the filter inductor voltage reference u*sLf from the supply voltage us s.
Since the aim in using the HPF is to improve the filtering performance with high order harmonics, the passive filter can be tuned to the frequency where the filtering performance of the active filter is impaired, i.e. close to 1 kHz. The filtering of the harmonics is now divided between the two filters: the APF focuses on the lower order harmonics, while the HPF filters harmonics remaining after active filtering. In conventional shunt APFs the filter inductor inductance Lf is a compromise between current control effectiveness and supply current switching ripple. In the hybrid filter the current control can be further improved by using a smaller filter inductor Lf. In a conventional shunt active filter this would raise the switching ripple in the supply current, but in the hybrid filter the ripple is filtered with the HPF.
The proposed method was tested in the laboratory with an active filter prototype shown in Fig 4. The passive high pass filter is not included in the figure. The prototype has been designed to compensate harmonic currents produced by a nonlinear load of 5 kVA nominal power (Us,LL= 400 V). The PWM bridge was built using 1200 V, 40 A IGBTs. The other prototype parameters are shown in Table I. The resonance frequency of the passive high pass filter was chosen to be about 850 Hz. The control system was implemented using a Motorola MPC555 microcontroller. This is a 32-bit single-chip microcontroller, whose main features are: 448 Kbytes on-chip FLASH EEPROM, 26 Kbytes on-chip RAM, two 8- or 10-bit A/D-converters with 41 input channels each, Modular I/O system (MIOS), two Time Processing Units (TPU) and two Serial Communication Interfaces (SCI), 0.1 Ã‚Âµs floating-point multiply and 0.25 Ã‚Âµs floating-point divide (at 40 MHz clock frequency).
The time scheduling of the control tasks is based on the interrupts made by the timer unit. The interrupts are generated twice in every modulation period. The modulation frequency is set at 10 kHz, resulting in an interrupt rate of 50 Ã‚Âµs. The software procedures are divided into three different priority levels. At the highest level there are protection and time-critical processes like feed forward and closed loop current controls, control delay compensation and the modulator updating. At the second level there is the dc link voltage control that is performed every 75th interrupt time after high priority level tasks. At the lowest priority there are the system synchronization and the user interface updating. These are done at the main program level.
Hybrid Filter Parameters
Supply phase voltage Us 230V
Supply frequency fs 50Hz
Filter inductor Lf 2.5mH
Dc-link capacitor Cf 1.1mF
Smoothing inductor Lsmooth 2.3mH
Switching frequency fsw 10KHz
Passive filter inductor Lhp 2.3mH
Passive filter capacitor Chp 15Ã‚ÂµF
Passive filter resistor Rhp 3O
Supply filter inductor Ls 1mH
Figure 4. Prototype. 1 drivers, 2 dc link capacitors, 3 IGBT bridge, 4
microcontroller, 5 supply filter Ls, 6 filter inductor Lf, 7.current measuring
To verify the proposed system filtering performance, two different loads were used in the measurements. First, a diode rectifier that supplied RL load produced the harmonics to be
compensated. The load of the diode rectifier consisted of a 10 mH inductance and 64 resistance connected in series. In the other case the harmonic producing load was a diode bridge that supplied an RC load where a 64 resistance was connected in parallel with a 1 mF capacitance. The measured phase-a load current waveform in the case of the RL load is shown in Fig. 5 and the RC load in Figs. 7. The waveforms in Figs. 7a and b are very similar, but there is a slight difference depending on the active filter topology used, to be explained later. Figs. 6a â€œ d present experimental results with APFs and Table II the harmonic content of the waveforms in the case of the RL load. The active filter dc link voltage reference has been set at 680 V. In Fig. 6a a shunt APF with a conventional, load current feedforward connection based control system has been used. In the case where the control delay compensation method has been applied to the control system of the shunt active filter, the supply current waveform is presented in Fig.
6b. It can be seen in Table II that the method improves the filtering performance of the low order harmonics. In Figs. 6a and b the filter inductor inductance Lf of the shunt APF was 5
mH. The supply current waveform with passive HPF parallel to active filter is presented in Fig. 6c. Now the switching ripple is filtered from the supply current, but clear glitches caused by
the control delay can be seen in the waveform. Fig. 6d presents the supply current waveform with the proposed method. With this system the benefits of both the computational control delay
compensation method and the shunt active filter with passive HPF are achieved. The system effectively filters harmonics under 2 kHz but also higher frequencies. With the proposed
method the total harmonic distortion is reduced from 27.58 % to 2.29 %. Figs. 7a â€œ b present the measured load current waveforms when the diode bridge supplies an RC load. Fig. 7a corresponds to the situation when the shunt APF is used and Fig. 7b when the hybrid filter is used. Because of the supply inductor in the hybrid filter topology, the active filter operation also affects the voltage ul seen by the load. This reflects on the currents drawn by the diode bridge, since the current drawn by the capacitor depends on the voltages seen by the bridge. This is why the load currents have different harmonic content in Table III. For the active filter to be able to produce compensating currents needed, the dc link voltage is controlled to be 750 V. The supply current waveforms corresponding to Figs. 7a â€œ b are shown in Figs. 8a â€œ b respectively. The harmonic content of the waveforms is shown in Table III. In Fig. 8a the shunt APF with the control delay compensation method is used and the proposed hybrid filter system is used in Fig. 8b. It can be seen in the table that the load current to be compensated also contains harmonics of the third order. They are caused by the distorted supply voltages supplying the three phase diode bridge. The fifth and seventh harmonics in the supply voltages give rise to e.g. a third harmonic component in the diode bridge currents. Table III shows that in the supply current THD2 kHz there is almost no difference between the methods but using the proposed method the THD20 kHz is about 1.5 percentage units smaller than with the shunt APF.
The performance of the proposed method in the transient state of the load is shown in Figs. 9 â€œ 10. In Figs. 9 an RL type load and in Figs. 10 an RC type load is used. In the figures the load resistance changes stepwise from 64 to 193 and back to 64 . As can be seen, the harmonics are also effectively filtered and the supply current is kept sinusoidal under transient
state of the load.
Figure 5. Measured phase-a load current waveform, using a three
phase diode bridge supplying an RL load.
Figure 6. Measured phase-a supply current waveforms in the case of the RL load current presented in Fig. 5, using a) the shunt APF with a conventional control system b) the shunt APF with the control delay compensation method c) the shunt APF with the HPF and d) the proposed method.
Figure 7. Measured phase-a load current waveforms, using a three phase diode bridge supplying an RC load. a) Shunt APF in operation. b) Hybrid filter inoperation
Figure 8. Measured phase-a supply current waveforms in the case of RC load current presented in Fig. 7 using. a) the shunt APF with the control delaycompensation method. b) the proposed method.
Figure 9. Measured phase-a current waveforms using the proposed method in the case of step chage in the RL load. a) Load current. b) Supply current.
Figure 10. Measured phase-a current waveforms using the proposed method in the case of step chage in the RC load. a) Load current. b) Supply current
[%] Shunt APF
[%] Shunt APF
5 23.07 2.63 1.39 2.19 0.91
7 9.88 1.76 0.71 0.33 0.38
11 7.95 2.24 0.40 1.44 0.07
13 5.21 1.44 0.45 0.60 0.25
17 4.13 1.69 0.60 1.30 0.10
19 3.05 1.33 0.71 0.79 0.24
23 2.27 1.10 0.93 0.88 0.67
25 1.83 1.12 0.85 0.49 0.62
29 1.26 1.01 0.76 0.94 0.82
31 1.10 0.95 0.91 0.51 0.85
35 0.70 0.65 0.56 0.63 0.38
37 0.63 0.60 0.75 0.33 0.58
THD2 kHz 27.55 5.18 2.74 3.58 2.07
THD20 kHz 27.58 6.12 5.91 3.70 2.29
TABLE II HARMONIC CURRENT COMPONENTS
n RC Load
[%] Shunt APF
[%] RC Load
3 9.22 1.11 5.52 0.58
5 58.31 1.99 56.38 1.07
7 33.00 2.12 30.70 1.22
9 1.89 0.41 1.10 0.04
11 18.46 0.49 7.98 0.18
13 5.13 1.09 4.98 0.83
15 1.29 0.31 0.71 0.05
17 3.05 0.47 2.88 0.11
19 2.29 1.023 2.11 0.94
21 0.94 0.49 0.63 0.15
23 1.94 0.94 2.24 1.09
25 1.38 0.80 1.36 0.96
27 0.71 0.57 0.36 0.41
29 1.23 1.03 1.11 1.39
31 0.83 0.85 0.96 1.37
33 0.50 0.49 0.29 0.56
35 0.78 0.84 0.69 1.63
37 0.62 0.90 0.62 1.45
39 0.41 0.58 0.27 0.88
THD2 kHz 68.57 4.31 65.31 4.07
THD20 kHz 68.58 6.52 65.32 5.05
TABLE III HARMONIC CURRENT COMPONENTS
To study the power losses caused by the system examined, power measurements were performed. The aim was to compare the efficiency of the proposed hybrid filter system to the shunt APF controlled using the control delay compensation method. Both systems were controlled so that the power factor of the supply was kept at one. In the measurements the assumption made was that the load is equal in every phase. This why only phase-a supply and load voltages and currents were measured. That is, with the shunt APF the measured quantities were usa, isa and ila and with the hybrid filter they were usa, isa, ula and ila. The measurement results are shown in Table IV. Increased compensatory current, when the harmonics of a diode bridge with an RC type load are compensated, causes increase in the power losses of both filter systems. It can be seen in Tables II and III that capacitive type load current has considerably greater harmonics of the orders 3, 5 and 7 than the inductive load type. To compensate them, the active filter has to generate harmonics of equal magnitude. This leads to an increase in the
power losses in the filter inductor Lf. The dc link voltage used has also an effect on the increase in power losses in both systems. To compensate the harmonics of the RC type load, the systems
used an active filter dc link voltage of 750 V. This is greater than the voltage used in the case of RL type load (680 V). This is why switching losses in the IGBT bridge increase. Table IV demonstrates that even the hybrid filter is a more complicated system with more components, it has almost the same efficiency as the conventional shunt active filter. When the harmonics of the diode bridge with the RL load are compensated, the shunt active filter has slightly lower losses than the hybrid filter, but with the RC type load the efficiencies of the systems are equal. Although the power losses are the same as in the shunt APF, the hybrid filter achieves better filtering result, as can be seen in Table III. With the RL load the efficiency of the hybrid system is lower, but it has a better overall harmonic filtering (Table II).
with RL load (%) RC load(%)
Method 95.3 4.7 90.9 .9
without Control 99.1 .9 99 1
APF + CDC 96 4 90.9 9.1
TABLE IV. FILTER SYSTEM EFFICIENCY
In this paper a method to achieve wideband harmonic compensation with a voltage-source hybrid active power filter was presented. The proposed method combines two methods to improve the performance of the shunt active filter: the computational control delay compensation was used to improve the filtering with low order harmonics while the higher frequencies were filtered with a passive high pass filter. The passive filter size was kept quite small, utilizing the whole filtering band of the active filter and tuning the passive filter resonance frequency where the active filter performance is degraded. The hybrid filter performance was examined with different kinds of load through measurements on a prototype in a laboratory and the results were compared to other filtering methods. The results prove the effectiveness of the proposed system for filtering the harmonics. The efficiency of the proposed system was compared to the shunt active filter and it was seen that there is only a small difference when harmonics of an RL type load are compensated, but with an RC type load the efficiencies are equal.
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